SRPP Demystified

SRPP

There is considerable misunderstanding and mystification regarding this particular circuit, including even what it's called. "SRPP" seems to mean: Shunt Regulated Push Pull, although you see different words assigned to the acronym.

As for what, exactly, this thingy is, it should be immediately recognizable to anyone with a solid state design background: it's an active pull-up/active pull-down circuit. (It greatly resembles the output stage of the TTL family.) The main difference is that it is quasi-complimentary by necessity since there is no such thing as a "P-Channel" VT. This means it is, indeed, push-pull by definition. The upper triode acts to source current to the load, and the lower triode sinks current from the load. As with any push-pull topology, it reduces distortion by nulling even order harmonics.

The big point of departure from solid state is that the SRPP is balanced for three load conditions only: a dead short (makes both triodes into grounded cathode stages -- not very useful) an open circuit (or at least a very high load impedance -- if there is just one path for the current, equal currents must flow through both triodes) and the one impedance for which it was designed. For any other load conditions, the SRPP goes out of balance, and distortion rises, more or less, rapidly. This is OK, considering the purpose for which the SRPP was originated: a line driver. As a line driver, it operates into the characteristic impedance of the T-line.

As an audio circuit, this leaves a lot to be desired. If it works into Class A*₁ grids, then it's OK since that's nearly an open circuit. The trouble starts when the driven control grids are driven positive, and draw current. Under grid current conditions, the resulting impedance is neither constant nor linear. This is not the type of load an SRPP wants to see. Including such a driver will lead to poor clipping behaviour. Unfortunately, it is grid current conditions where you'd like to include active pull-up to not only supply that grid current, but supply it from a Lo-Z source to minimize distortion.

You also see SRPPs used as audio finals. This, too, is not the place for this since any speaker represents anything but a constant load. Speaker impedance varies not only in magnitude, but also phase angle. This will play hell with an SRPP, as it will be operating off its optimum load impedance almost all of the time. That will generate lots of avoidable distortion which will require that much more NFB to correct. This is not what you want in good open loop design.

As for how the SRPP develops voltage gain, consider the R_P. The far end of that plate load resistor has a cathode follower sitting on top of it. This means that the voltage across that resistor is much less than it would be if connected directly to the DC rail. The effective AC resistance is much higher than its DC resistance. This gives the lower triode more voltage gain that it would otherwise have. The next question becomes: can we make that resistor even larger to increase voltage gain? If you break the DC coupling, the answer is "yes". This gives us the variation called a "Mu stage", so called since its voltage gain can approach the amplification factor (μ) of the lower triode.

Mu Stage

The AC coupling allows for a larger R_P since it is no longer doing double duty as a cathode bias resistor. The mu stage no longer has any pretension for being a balanced topology. It is designed for large voltage gains. Is this a useful topology? It was back in "the day", however, it is obsolete and should not be used. Today, we have solid state devices which can operate as excellent CCSs. You will do much better loading a triode with a constant current, as this gives a horizontal loadline that both maximizes output swing, voltage gain, and minimizes harmonic distortion. Back in "the day" we didn't have the ICs, BJTs or MOSFETs that could have made for decent CCSs. Before then, your only other recourse was to use a pentode as a CCS. That would give you something quite mu stage-ish anyway.

If you need lots of gain, then use a solid state CCS. These old fashioned circuits serve no purpose these days, other than nostalgia appeal, or audiophool trendiness on the part of those who like "exotic" circuits for the exoticness.

Cathode Follower

 

AC Coupled CF

In this schemo, R_K is the normal cathode bias resistor. R_L represents the tail load in parallel with the load impedance. R_G is the control grid DC return. The CF gives excellent high frequency performance since Miller Effect is absent, and the C_GK sees very little current since the grid and cathode are always at nearly the same potential. This makes the C_GK effectively smaller than its static value. The main component of input capacitance will be the reverse transfer capacitance: C_GP. With small signal triodes, it is easy to present a Hi-Z, Lo-C load to the driving stage. This isn't just helpful at RF.

This is another circuit which has lately come under unjustified criticism within certain audiophile circles. Much of this is unjustified on the basis that the CF is a negative feedback circuit. This view that all NFB is all bad does have a basis in fact. It has been all too common to use NFB to cover up for poor open loop designs. If your open loop design is poor, just pour on the NFB to force "the numbers" to look good. Sure, you can sweep your mistakes under the carpet that way, but you will also sweep away much of the vitality of the music. However, this is a misuse and abuse of NFB. The blame properly belongs to these lazy designers who can't be bothered to correct their open loop designs.

Another big part of the problem lies with the nature of the cathode follower itself. Yes, it can present a low impedance source, but only to a high impedance load. There is a big difference between source impedance and load impedance. For example, a 6C4 small signal triode could be used to implement a cathode follower. If you implemented it thusly:

V_PP= 330V_DC
V_PKQ= 140V_DC
I_PKQ= 4.0mA
R_tail= 47K

r_p= 10.5K
g_m= 1.4mA/V

You could easily calculate a Z_o= ~647Ω Given that Z_o, you might think you could drive a set of 600Ω headphones with this cathode follower. However, you would be quite wrong about that. Your undistorted power output will be just under 10mW as you'll only be able to swing just 3.4V_p into that load. So what happened? When you connected a 600Ω load across the 47K tail, you killed most of your gain by making a nearly vertical loadline. Less open loop gain means less effective NFB. If it can not drive a 600Ω load connected to the plate it can not drive that load any better if you connect it to the cathode.

So what is it good for? This cathode follower would be ideal for isolating a grounded cathode gain stage from, let's say, a tone stack. The GC amp will have a very large Z_o. You could incorporate that source impedance into the resistances of the tone stack, to be sure. However, tube characteristics vary considerably with the manufacturer, models within brands, with age. The output impedance probably won't stay put, and if it varies, the poles 'n' zeros of your tone stack will change with it. Isolating the tone stack with a cathode follower representing an insignificant portion of the tone stack resistances will prevent this. It will also allow for more resaonable values of resistance and capacitance in the implementation.

Cathode followers can also be used as active pull-up circuits to drive the control grids of audio finals. Even if you stay with Class A*₁ operation, the input capacitances (C_GK + C_Miller + C_stray) are still going to require current to charge. If the current sourcing capability isn't there, then you will run into slew limiting at the higher audio frequencies. That sounds nasty. The CF can supply enough current to prevent this from happening, especially if you follow the "Rule of Five" from solid state practice: make the Q-Point current of your CF at least five times greater than your anticipated peak current.

Don't ever forget: the vacuum tube itself neither knows nor cares whether the load is connected between the plate and the positive rail, or if it's connected between the cathode and DC ground. It's always the same loadline, the same load resistance. You don't gain anything by trying to force the device into being something it will never be: a high current, low voltage device. A CF is not a magical power gain stage.

Attempting to use it otherwise will lead to degraded sonic performance. If you have a bad-sounding CF, blame the designer, not the topology.

As with any other audio subsystem, if the CF is designed properly, and used within its limitations, it is the most sonically transparent audio subsystem. If designed badly, it will sound bad. It's as simple as that.

Demystifying the Cascode

Basic Cascode

This is what the schemo of the cascode looks like. R_K serves to establish the Q-Point bias for the lower triode, and R_G is its DC grid return. R_P is the passive plate load. The voltage divider connected to the grid of the upper triode establishes its Q-Point bias.

So why would you want to do this? What you have here is a cascade of a grounded cathode stage driving a grounded grid stage. The GC topology has the advantage of a Hi-Z input. However, its high frequency performance is impacted by a high C_Miller that only grows worse with increasing voltage gain.

The GG topology avoids C_Miller for excellent high frequency performance, but it suffers from a Lo-Z input. It's not very often that a Lo-Z input is desirable. However, you can combine the two in a manner that work together. The Lo-Z of the GG stage loads down the plate of the GC stage, reducing its gain greatly. The lion's share of voltage gain comes from the GG second stage. Reducing the gain reduces C_Miller to manageable levels while preserving the Hi-Z input. This gives the cascode a characteristic more like that of a small signal tetrode, with its reduced C_Miller, high voltage gain, less the screen grid "kinks", and partition noise. Unlike a tetrode, the V_GK of the upper triode remains negative, and so you also don't get the partition noise that tetrode screen grids produce. That's where the name comes from: a contraction of "cascade" and "tetrode".

It is for this reason that the cascode is frequently cited as a VHF small signal amplifier. As for what this means for audio amplification, the reduced C_Miller is helpful since a volume pot with a high resistance won't produce the roll-off that bothers designs that place the high gain triode stage up front. The cascode, having higher gain than a single triode gain stage also looked quite useful. As an LTP phase splitter, the cascode could give enough gain to eliminate a second gain stage.

As for sonic performance, there is very little information concerning this. Most of the information I could find related to the design of guitar amps where voltage gain was the emphasis, as distortion is much less of a consideration in such designs. Cascoded LTPs are used quite frequently in solid state designs. However, I could not come up with any examples of hollow state designs that used this topology. Was that just because it was "weird", or required another dual triode, or had this been tried and found to be sonically inferior?

This was another case of try it to find out. As for VT selections, the 6SN7 didn't provide enough gain, and the 12AT7 suffered a fast g_m roll-off with decreasing plate current. Digging into the RCA Receiving Tube Manual (RC-30), I came up with the 6BQ7A -- a dual triode designed specifically for cascoding and operation up to 300MHz. Its μ= 38 is considerably higher than that of the 6SN7, but is still reasonable for this particular design. The main problem with the 6BQ7 is that it isn't an audio tube, has no audio use specified in the spec sheet, and has a very peculiar plate characteristic (undocumented variable-μ feature?). Finding a good audio loadline is not so easy, and this type likes to see a V_PK that's higher than usual. Still, it looked doable, and since this is a balanced topology, it should greatly reduce harmonic distortion if most of that is h₂.

Cascoded LTP Design

There are a couple of bug-a-boos with this topology. The output voltage swing is rather small for the DC rail voltage, the PSRR is less than that of most triode-only gain stages, and the output impedance is very high. This last feature is desirable in an RF amp since that means less loading of LC tuners, and higher loaded Q's. For an audio amp that has to operate over a rather large range of frequencies, it's not such a good thing. A Hi-Z output will interact poorly with the input capacitance of a subsequent stage or other load. To prevent premature roll-off, it is necessary to operate into a Hi-Z, Lo-C load.

The design also uses an active (CCS) tail load. The CCS presents a very high impedance to the junction of the two cathodes. This greatly improves both the phase-to-phase AC balance, and also equalizes the harmonic distortion between phases. It is this distortion imbalance that gives a great many phase splitters inferior sonics. The CCS is a cascode of BJTs. This gives better performance than would the more traditional small signal pentode. The BJT, having much greater gain, gives a higher tail impedance, and a more nearly constant current, with a lower negative rail voltage. This is one area where solid state really is better.

O'scoping the output of the cascode directly resulted in poor square waves with severely tilted tops, indicating a loss of high frequencies. The -3db_v point was barely 20KHz. That's positively horrible! However these were artifacts of the C_i of the o'scope probe, cable, and vertical deflection amp. So this meant it required a "friendlier" load. For that, a cathode follower works nicely. A CF stage has the high input impedance, and the low input capacitance since there is no Miller Effect. Since the voltage at the cathode "follows" the grid, there is very little current through C_GK, so this capacitance all but disappears. When operated into a cathode follower, the true picture emerged: nice flat square waves to 10KHz, a -3db_v point of 117KHz, more than enough for an audio amp.

So how did it sound? In a word: excellent. There were no pentode-like artifacts at all, and the sonics were identical to what a good triode LTP would produce. It beats all the paraphase splitters in their various iterations. This definitely should see much wider use than it does.

Welcome to Dolphin Hollow State Labs

 

For audio amplification, the first active device remains the best active device: the triode vacuum tube. The triode is unique among the active devices: vacuum tube pentodes and transistors of all sorts. With these devices, the plate/collector/drain current is largely independent of the voltage across the device. This property makes them excellent approximations of an ideal current source.

With triodes, however, this is not the case. The plate current can vary with the plate-to-cathode voltage. This is the meaning of "amplification factor" --

μ= ΔV_pk / ΔV_gk (at Ip= constant)

For most triodes, the amplification factor will usually be between 10 to 100. For power triodes, this can be smaller (audio power finals, vertical deflection power amps, series pass regulators) or higher ("zero bias" RF finals). Amplification factor is largely meaningless for other active devices since it's so difficult to measure directly.

The triode, like every other active device, has an inherent degeneration when operated as a grounded cathode amplifier. This being the cathode resistance: r_k= ~1 / g_m. This resistance acts in precisely the same manner as if it were an unbypassed resistor soldered into the circuit. Just because you don't see it doesn't mean it's not there.

There is another feedback mechanism at work as well. When the signal pulls the V_gk less negative, the plate current increases, and with it, the voltage drop across the plate load. This results in a decreasing V_pk. As this voltage decreases, it tries to pull the plate current lower. Since V_gk and V_pk are pulling in the opposite direction, this is negative feedback by definition.

It is this additional source of NFB that serves to correct for harmonic distortion to a greater extent than you will see with other active devices. What harmonic distortion remains is mainly the second harmonic, h₂. Sonically, h₂ is the least detrimental. It is this h₂ that lends to the so-called "tube sound", described as "warm", "rich", "full", etc. More h₂ is described as "dark". No distortion would be the best, but the perfect amplifier, like the perfect lens for telescopes, has yet to be invented. The question becomes how to minimize those defects which you will never completely eliminate.

Of course, not all triodes are equally good sounding. As far as the effectiveness of plate current control by plate voltage, this is measured by the term: plate resistance: r_p. A good many high-μ triodes attain that large amplification factor by driving up the plate resistance. For the 12AX7, the spec sheet gives: r_p= 80K (nominal). This is comparable with the r_p's of small signal pentodes. Is it any wonder why the 'AX7 tends to sound like a small signal pentode?

Even if the r_p isn't excessive, some triodes just won't perform all that well for audio. This being caused by excessive variation in g_m with plate current variations. For all active devices, amplification tends to increase with increasing current. If the positive going half cycle of a sine wave receives more amplification than the negative half cycle, then the two half cycles hit different peak voltages. That's not a sine wave any more, and something has been added. Since this is asymmetrical distortion, it is even order, and mainly h₂. Types such as the 12AV7 tend to produce much more of this distortion than you'd like to see. (Although it just might be useful for audio effects.)

Nothing sounds better. It is for this reason that this "obsolete" device is still around over a century after its invention. This has been known at least since the early-1950s when the quest for a solid state device that could match the distortion performance of the VT triode began. So far, no one has found such a device.